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MIC3263
Six-Channel WLED Driver for Backlighting Applications with Flicker-Free Dimming
General Description
The MIC3263 is a high-efficiency Pulse Width Modulation (PWM) boost switching regulator that is optimized for constant-current WLED driver backlighting applications. The MIC3263 drives six channels of up to ten WLEDs per channel. Each channel is matched in current to within 3% for constant brightness across the screen and can be programmed from 15mA to 30mA. The MIC3263 provides a very flexible dimming control scheme with better accuracy and noise immunity. The dimming frequency can be set to any value between 100Hz and 20kHz by an external resistor. The dimming ratio is determined by the duty cycle of a dimming ratio control input signal and can be set to one of 16 levels with a minimum ratio of 1%.The LED dimming current is set by an external resistor to allow programming of LED current between 15mA and 30mA. The dimming ratio of the MIC3263 is fixed to 16 log levels to better match the sensitivity of the human eye. Each of the dimming levels has hysteresis to avoid skipping between levels and allow for high noise immunity. The MIC3263 has a programmable PWM switching frequency from 400 KHz to 1.8 MHz to allow small inductor sizes. The 6V to 40V wide input voltage range of MIC3263 allows direct operation from 6V or high cell count Li-Ion batteries commonly found in notebook computers. The MIC3263 is available in a low-profile 24-pin 4mm x 4mm MLF(R) package and has a junction temperature range of -40C to +125C. Data sheets and support documentation can be found on Micrel's web site at: www.micrel.com.
Features
* * * * * * * * * * * * * * * * 6V to 40V wide input voltage range Drives 6 channels of up to 10 white LEDs Programmable WLED current from 15mA to 30mA Highly reliable operation with open and short LEDs Accurate 16 dimming log levels sets the dimming ratio from 1% to 100% Flicker-Free Dimming filters the jitter from the dimming control input signal and eliminates dimming flicker Allows external dimming control Accurate LED channel current matching 3% Accurate initial LED current setting 2% Programmable switching frequency from 400kHz to 1.8MHz High efficiency up to 90% Low (<40A) shutdown current over temperature Over temperature protection Programmable over-voltage protection -40C to +125C junction temperature range Available in 24-pin 4mm x 4mm MLF(R) package
Applications
* * * * * * * * White LED driver for backlighting Notebooks LCD Panels and Monitors Multimedia players Navigation equipment Gaming systems Video poker Slot machines
_________________________________________________________________________________________________________________________
MLF is a registered trademark of Micrel, Inc. Micrel Inc. * 2180 Fortune Drive * San Jose, CA 95131 * USA * tel +1 (408) 944-0800 * fax + 1 (408) 474-1000 * http://www.micrel.com
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MIC3263
Typical Application
MIC3263 Typical Application Schematic
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Ordering Information
Part Number MIC3263YML
Note: 1. Other Voltage available. Contact Micrel for detail
Junction Temperature Range(1) -40 to +125C
Package 24-Pin 4mm x 4mm MLF(R)
Lead Finish Pb-Free
Pin Configuration
24-Pin 4mm x 4mm MLF(R)
Pin Description
Pin Number 1 2 3 4 5 Pin Name FSW RSLP OVPS OVP MODE Pin Function Booster Switching Frequency: Connect a resistor-to-GND to set the switching frequency from 400kHz to 1.8MHz. Slope Compensation Adjustment Resistor. OVP and FB voltage divider virtual ground. Overvoltage Protection Input. This is also the FB voltage for the error amp in the Boost stage. Select a dimming frequency range: 0V for 100Hz to 2kHz and VDD for 1.5kHz to 20kHz. If DFS is connected to VDD, MODE pin is used for an external dimming pulse input. 6 DFS Set a dimming frequency from 100Hz to 20kHz through an external resistor and MODE. Requires a series RC for stability. If DFS is connected to VDD, an external dimming pulse can be applied to the MODE pin. 7 8 COMP DRC Loop Compensation connect R and C-to-GND. Dimming Ratio Control Pulse: Its duty cycle is converted to one of 16 dimming levels. The duty-cycle difference between two adjacent levels is 6.25%. And about 2% duty-cycle hysteresis exists between two adjacent levels to eliminate dimming flicker. DRC can be from 100Hz to 40kHz. 9 CINT Integration Cap: Use a 0.01F for 2kHz to 20kHz and 0.1F for 100Hz 2kHz.
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Pin Description (Continued)
Pin Number 10 Pin Name ISET Pin Function LED Dimming Current Set: Connect a resistor-to-GND to set the dimming current from 15mA to 30mA. Use 2k for 30mA, and 3k for 20mA. 11 12 13, 14, 15, 16, 17, 18 19 20 21 22 23 24 CRV AGND IO1 IO6 NC PGND VSW EN VIN VDD EP Capacitor reference voltage: Connect a 2.2F capacitor-to-GND. Analog signal Ground. LED Channel Current Sinker: Connect the cathode of each channel of LEDs to one current sinker. No Connect. Power Ground. Switch Node: Internal power NPN collector. Enable Pin: Connect HIGH or LOW; do not float. Supply: 6V to 40V. Output of internal LDO: Connect a 10F capacitor-to-GND. Connect to PGND
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Absolute Maximum Ratings(1)
Supply Voltage (VIN), Enable (VEN)...............................+42V Switch Voltage (VSW)..................................... -0.3V to +42V Regulated Voltage (VDD) ................................. -0.3V to +6V Over-Voltage Protection (VOVP) .................... -0.3V to +42V Switch Voltage (VOVPS) ................................. -0.3V to +42V DFS Voltage (VDFS) ............................ -0.3V to (VDD + 0.3V) RSLP (VRSLP) ...................................... -0.3V to (VDD + 0.3V) MODE Voltage (VMODE) ...................... -0.3V to (VDD + 0.3V) FSW Voltage (VFSW) ........................... -0.3V to (VDD + 0.3V) DRC Voltage (VDRC) ........................... -0.3V to (VDD + 0.3V) CRV Voltage (VCRV)............................ -0.3V to (VDD + 0.3V) CINT Voltage (VCINT) .......................... -0.3V to (VDD + 0.3V) ISET Voltage (VISET) ........................... -0.3V to (VDD + 0.3V) Comp Voltage (VCOMP)........................ -0.3V to (VDD + 0.3V) IO1-IO6 Voltage (VIO1-IO6) ............................. -0.3V to +42V AGND to PGND ........................................... -0.3V to +0.3V Lead Temperature (soldering, 10 20s) ................... 260C Storage Temperature (TS)......................... -65C to +150C ESD Rating(3) ............................................................... 1.5kV
Operating Ratings(2)
Supply Voltage (VIN)......................................... +6V to +40V Enable (VEN) ..........................................................0 to +40V MODE (VMODE)......................................................0 to +5.5V DFS (VDFS)............................................................ 0 to +5.5V DRC (VDRC)...........................................................0 to +5.5V Junction Temperature (TJ) ........................ -40C to +125C Junction Thermal Resistance 24-Pin MLF(R) (JA) .............................................43C/W
Electrical Characteristics(4)
VIN = 12V; L = 22H, COUT =10F,TA = 25C, BOLD values indicate -40C TJ +125C, unless noted. Symbol VIN VIN IVIN VDDREG ISD IO1 IO6 VOS Parameter Supply Voltage Range Supply Voltage Range Quiescent Current VDD Regulation Shutdown Current (DC Pin Low) Minimum IO (1-6) Voltage for operation to Sink 30mA Maximum Output Voltage Overshoot when Current Sources are OFF in PWM Dim Mode Channel Current Matching Initial Current Setting Accuracy PWM Dimming Frequency Adjust Range Condition 30mA 8 LEDs/Channel, All six Channels 30mA 6 LEDs/Channel, All six Channels Not Switching, VOVP = 4V VIN = 6V to 40V, IDD = 0mA to 6mA VEN = 0V Voltage on IO (1-6) if Only One Channel is Used and ISET = 30mA 22H, 10F 4.5 Min 8 6 6.5 5 6.5 1.2 3 Typ Max 40 40 10 5.5 20 A V % Units V V mA
Current Control
ILEDMATCH ILEDSET FDIMR
ILED = 30mA and Dimming Ratio = 100% VIO = 1.2V on All Channels RSET = 2k ILED = 30mA MODE = 0V, RDFS = 400k, Frequency = 100Hz MODE = 0V, RDFS = 32k, Frequency = 1.2kHz MODE = VDD, RDFS = 400k, Frequency = 1.6kHz MODE = VDD, RDFS = 32k, Frequency = 20kHz
-3 -2 -3 0.1
0 0
+3 +2 +3 20
% % kHz
Notes: 1. Exceeding the absolute maximum rating may damage the device. 2. The device is not guaranteed to function outside its operating rating. 3. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF. 4. Specification for packaged product only.
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Electrical Characteristics(4) (Continued)
VIN = 12V; L = 22H, COUT =10F,TA = 25C, BOLD values indicate -40C TA +125C, unless noted. Symbol FDIMA FDRC VPWM VEN IEN Boost Converter DMAX ISW VSW ISW N FSW fSW VOVP TSD Maximum Duty Cycle Switch Current Limit Equivalent Switch VCE(ON) Switch Leakage Current Efficiency Oscillator Frequency Range Oscillator Frequency Overvoltage Protection Thermal Shutdown VIN = 6V to 20V, Guaranteed by Design VIN = 12V, ISW = 1.0A VIN = 0V, VIN = 40V VIN = 12V, Load = 6 Channels of 8 LEDs at 20mA with 3.6V per LED, Frequency = 400kHz Frequency Setting Range RFSW = 160k Comparators OVP Pin to 2.36V Temperature Rising Hysteresis 0.4 0.96 90 1.6 2.4 0.3 0.01 90 1.2 1.2 2.36 160 20 1.8 1.44 20 % A V A % MHz MHz V C Parameter PWM Dimming Frequency Accuracy DRC Input Range DRC Pin Thresholds EN Pin Thresholds Enable Pin Current Turn on Turn off Turn on Turn off 40 1.3 0.4 60 A Condition FDIM = 100Hz to 2kHz; MODE = 0 FDIM = 1.6kHz to 20kHz; MODE = VDD Min -20 -20 0.1 1.3 0.4 V Typ 0 0 Max +20 +20 40 Units % kHz V
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Typical Characteristics
Dimming Efficiency vs. Input Voltage
100% PWMD 72% PWMD 52% PWMD
ILEDs vs. Input Voltage
30.5 30.4 30.3 30.2 ILED (mA)
Ch5 Ch4 Ch1
95 90 EFFICIENCY (%) 85
2.0 1.5 ILED (% Change) 1.0 0.5 0.0 -0.5 -1.0 -1.5
% Change in ILEDs vs. Input Voltage
Ch3
30.1 30.0 29.9 29.8 29.7 29.6 29.5
Ch6 Ch3 Ch2
Ch2 Ch1
37% PWMD
80
27% PWMD 19% PWMD
Ch5 Ch6 Ch4
75 70 6 11 16 INPUT VOLTAGE (V)
5
10
15
20
5
10
15
20
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
30.4 30.2
ILED @30mA vs. Temperature
Ch2 Ch3 Ch6
1.00 0.80 0.60
% Change in ILED@30mA vs. Temperature
Ch1 Ch5
15.30 15.25 15.20
ILED @15mA vs. Temperature
Ch6 Ch3
30.0
ILED (% CHANGE)
0.40 0.20 0.00 -0.20 -0.40 -0.60 -0.80 -1.00
Ch3
ILED (mA)
29.8 29.6 29.4 29.2 29.0 -40 -20 0 20 40 60 80 100 120
Ch1 Ch4 Ch5
ILED (mA)
Ch4
15.15 15.10
Ch5 Ch1
Ch2
15.05
Ch6 Ch4
15.00
Ch2
14.95
-40 -20
0
20
40 60
80 100 120
-40
-20
0
20
40
60
80
100 120
TEMPERATURE (C)
TEMPERATURE (C)
TEMPERATURE (C)
0.80 0.60
% Change in ILED @15mA vs. Temperature
Ch3 Ch6 Ch2
VDD Voltage vs. Input Voltage
5.00 4.90 4.80 4.70 4.60 4.50
SWITCHING FREQUENCY (kHz)
950
Switching Frequency vs. Input Voltage
0.20 0.00 -0.20 -0.40 -0.60 -0.80 -1.00 -40 -20 0 20 40 60 80 100 120
Ch1 Ch5 Ch4
VDD VOLTAGE (V)
ILED (% CHANGE)
0.40
25C
940
930
920
25C
910
5
10
15
20
900 5 10 15 20 INPUT VOLTAGE (V)
TEMPERATURE (C)
INPUT VOLTAGE (V)
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Functional Characteristics
VOUT, VSW, ILED at 10% Dimming Dimming Transient Response
LED Ripple Current
Line Transient Response
Switching Waveform
Start Up
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Functional Characteristics (Continued)
ENABLE Start Up PWM Dimming
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Functional Diagram
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Functional Description
The MIC3263 is a six-channel LED driver. A constant output current converter is the preferred method for driving LEDs. The MIC3263 is specifically designed to operate as a constant-current LED driver to keep the current in all six channels constant. PWM dimming is employed in each channel. Each channel of LED current is individually and tightly regulated during each Duty Ratios (DR) on-time. During the DR off-time the LED current is turned off. The duty cycle of the DR pulse determines the brightness of the LEDs. The MIC3263 is designed to operate as a boost controller in which the output voltage is higher than the input voltage. This configuration allows for the design of multiple LEDs in series to help maintain color and brightness. During each DR pulse off-time the boost converter is turned off (not switching). The boost converter is on (switching) during each DR pulse on-time. The MIC3263 has a very-wide input voltage range of 6V and 40V to help accommodate for a diverse range of input voltage applications. In addition, the LED current can be programmed through the use of an external resistor (RISET). This provides design flexibility in adjusting the current for a particular application. The MIC3263 can control the brightness of the LEDs via its PWM dimming capability. Applying a PWM dimming signal (up to 40kHz) to the DRC pin allows for control of the brightness of the LED. It has a boost stage that boosts the VIN to a high enough voltage to forward bias the LED channels. The MIC3263 is a constant current controller. The controller keeps the current in each of the six channels at a constant value. Each channel has an independent current regulator in series with each LED channel. The current in each channel is within 3% of the others. The MIC3263 uses three main control loops (Figure 1 control loops): 1) Current Amp loop (Fastest) 2) Booster loop (Fast) 3) Capacitor Reference Voltage (CRV) loop (Slow) The current Amplifier Loop is faster than the Boost Loop and CRV Loop. CRV is the reference voltage for the boost error amp.
Figure 1. Constant-Current Control Loops
Figure 2. Simplified Control Loop
The objective of these loops is to keep the LED current constant. The boost output voltage VOUT will vary when CRV changes. VOUT will be what it needs to be to keep ILEDs constant. The current amp loop is so fast the other loops can be viewed as static DC values. On a pulse to pulse basis the boost loop is fast enough that CRV is a constant value. The goal of the CRV loop is to keep the collector's voltage V(IO1-IO6) at or about 1.2V, thereby keeping the bipolar transistor in the linear region and also keep the power loss across the bipolar as low as possible. Keeping the bipolar in the linear region allows the current amp loop to be able to regulate the LED current.
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(R1+ R2) R2
MIC3263
VOUT = CRV x
The MIC3263 is designed for a wide input voltage range, from 6V to 40V. As a peak current-mode controller, the MIC3263 provides the benefits of superior line transient response as well as an easier to design compensation. MIC3263 provides several protection features, including: * * Current Limit (ILIMIT) Current sensing for over current and overload protection Over-Voltage Protection (OVP) output over-voltage protection to prevent operation above a safe upper limit The boost stage is on (switching) during a high DR pulse and is off (not switching) when the DR pulse is low.
*
Application Information
At Start Up At start up, a switch connects 1.8V to the CRV. The feedback resistor divider (R1 and R2) is calculated to achieve the approximate boost output voltage with a VCRV of 1.8V. Example: * * * * 8 LEDs at 3.5V each = 28V VIO = 1.2V VOUT = 29.3V estimate Set R divider to: R1 =150k R2 = 9.88k
The CRV control loop will charge/discharge CRV until the correct boost voltage appears at the output. Case 1 If 29.3V is too high to properly forward bias the LED channel at the ISET current level, then the current amp loop will decrease the drive to the bipolar transistor and V(IO) will increase and the CRV control loop will decrease CRV and the boost output voltage (VOUT) will decrease.
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Micrel, Inc. Case 2 If 29.3V is not high enough to properly forward bias the LED channel at the ISET current level, the current amp loop will drive the bipolar transistor harder and V(IO) will drop and the CRV control loop will increase CRV and the boost output voltage (VOUT) will increase. Internal Dimming Control In the internal dimming mode, the dimming is determined by the DFS and MODE pins. An external pulse is tied to the DRC pin. The duty cycle of the external pulse (pulse at DRC) is converted to one of 16 levels called Duty Ratios (DR) (see Table 2 for DR ratios). It is this internal pulse (DR) that is used to PWM dim the LEDs.
MIC3263
Faults
Open LED in Channel If any LED in a channel fails open, the voltage on the collector of the current amp transistor (IO1-IO6) will go low. The circuitry that monitors the IO pins will detect less than 0.5V and turn off base drive to the transistor. A flipflop latches the fault condition and a power down and power up sequence is required to reset that channel. Without base drive to the transistor, the channel of LEDs will turn off and a high impedance will be present at the collector (IO). The other five channels will continue operating normally. This fault sequence is identical if up to three LED channels fail open. If four channels fail open or short, then the remaining two LED channels will stay on and no more faults will be detected. Short LED in Channel If any LED in a channel fails shorted, the voltage on collector of the current amp transistor (IO1-IO6) will go high in voltage. If the circuitry that monitors the current amp bipolar transistor detects more than 7.5V at the collector (IO), then the base drive to the transistor will turn off. A flip-flop latches the fault condition. A power-down and power-up sequence is required to reset that channel. A channel can tolerate a two LED difference before a fault is detected. Without base drive to the transistor, the channel of LEDs will turn off and a high impedance is present at the collector (IO). The other five channels will continue operating normally. This fault sequence is identical if more than one LED channel fails open. If four channels fail open or short, then the remaining two LED channels will stay on and no more faults will be detected. Shorted Cathode (or IO Short) If the circuitry that monitors the current amp bipolar transistor detects less than 0.5V at the collector (IO), then the base drive to the transistor will turn off. A flip-flop latches the fault condition. A power-down and power-up sequence is required to reset that channel. Without base drive to the transistor, the channel of LEDs will turn off and a high impedance is present at the collector (IO). The other five channels will continue operating normally. This fault sequence is identical if more than one LED channel fails open. If four channels fail open or short, then the remaining two LED channels will stay on and no more faults will be detected.
Figure 3. Internal Dimming Control
External Dimming Control In external dimming mode, connect the DFS pin to VDD and apply a PWM dimming pulse to the MODE pin. The external pulse directly controls the LED current drivers (see Figure 4).
Figure 4. External Dimming Control
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Micrel, Inc. OVP An open LED in a channel will not trigger an OVP. OVP monitors the boost output voltage. If an open occurs on the load (all channels open) an OVP fault will trigger an overvoltage condition. When the OVP triggers, it turns off the boost and starts an OVP cycle. If one, two, or three channels open, they will not trigger an OVP. Four open channels will trigger an OVP fault and will cycle on and off at about 2Hz as long as there are four open channels. If one of the LED channels is reconnected (not open), then operation returns to normal for those three channels that are reconnected with out having to go through a power on reset. In the event of a load opening (four or more channels open) the following will occur: 1. VIO will drop below 1V 2. Charge pump will raise CRV during each DR pulse on-time 3. CRV will increase to 2.4V 4. When CRV reaches 2.4V the boost output maximum voltage will be; VOUT_MAX = 2.4* (R2+R2)/R1. 5. Feedback VOVP will reach 2.4V and the OVP comparator will trip and turn off the booster. 6. With the booster off, VOUT and VOVP will discharge. When feedback reduces to 1.7V the booster is turned back on. 7. The OVP circuit will switch 1.8V onto CRV 8. If the load is still open the cycle will continue.
MIC3263
Condition 1 LED Shorts 2 LEDs Short in Same Channel More Than 2 LEDs Short in Same Channels 1 LED Opens in Channel 1 2 or 3 Channels Open LEDs 4 or More Channels Open All Channels Open VOUT Shorted
Fault NO NO
Monitor IO > 1.2V 1.2 < IO < 7.5
Result All Channels On All Channels On 1 Channel Off; 5 Channels On 1 Channel Off; 5 Channels On 3 Channels Off; 3 Channels On 4 Channels Off; 2 Channels On OVP Triggered Output Current is Limited
YES
IO > 7.5V
YES
IO < 0.5V
YES
IO < 0.5V
YES
IO < 0.5V
YES
OVP Threshold Exceeded Current Limit Exceeded
YES
Table 1. Fault Summary
Power-On Sequence
VIN needs to be present before PWM pulses are applied to the DRC pin. Some channels may not turn on if the power up sequence isn't followed. This is because the circuits that monitor the IO pins may see transients during the turn on-time and may interpret voltage spikes during turn on as a fault, preventing that channel from turning on. When a channel is off, its IO pin is at high impedance. It is best to follow the sequence: 1. VIN 2. PWM dimming at DRC 3. Enable high
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MIC3263 Use the following equations to determine the value for RDFS: RDFS(k) = -20 x fDIM(kHz) + 432 (HF Mode) RDFS(k) = -335 x fDIM(kHz) + 433 (LF Mode) Example: For a dimming frequency of 10kHz, use the HF Mode: RDFS(k) = -20 x 10 + 432 = 232k in HF Mode For 1kHz, use LF Mode: RDFS(k) = -335 x 1 + 433 = 98k in LF Mode Use the closest standard value.
450 400 350
Pin Descriptions
FSW Sets the boost switching frequency. Connect a resistor from FSW to GND to set the switching frequency between 400kHz and 1.8MHz. Use the following equations to select RFSW: RFSW (k) 500 - 0.3 x fSW(kHz) RSLP The boost section is a peak current mode typology and needs slope compensation to eliminate sub-harmonic oscillation (see "Slope Compensation"). OVPS This is a virtual ground of the resistor divider feedback network in the boost stage. At turn on, a switch connects this node-to-ground. When the part is disabled the switch will open and disconnects the feedback resistor network from ground. This eliminates current draw from VIN by the boost resistor divider network. OVP This is the over-voltage protection monitor. Also this is the feedback signal that connects to the error amp input. MODE This selects the internal PWM dimming frequency range. When mode is low the PWM dimming frequency range is 100Hz to 2kHz. When mode is high the PWMD frequency range is 1.5kHz to 20kHz. Mode is high selects High Frequency (HF) mode; Mode is low selects Low Frequency (LF) mode. DFS DFS stands for Dimming Frequency Select. The dimming frequency of the LEDs is different than the input dimming frequency at the DRC input. The MIC3263 uses an internal dimming frequency. This internal dimming frequency is programmable by an external resistor to ground RDFS. For direct dimming control, connect DFS to VDD and use the MODE pin for the input dimming pulse. This method by passes the internal dimming control and allows for dimming control by the external PWM pulse. When using internal dimming the range is determined by the MODE pin and the actual frequency is determined by RDFS. Connect a resistor to ground to select a dimming frequency.
RDFS(in k) = -20*Dimming Frequency (in kHz) + 432
RDF S (k)
300 250 200 150 100 50 0 0 2.5 5 7.5 10 12.5 15 17.5 20 22.5
Dimming Frequency (kHz)
Figure 5. RDFS vs. Dimming Frequency in HF Mode
RDFS(k) = -335*Dim m ing Frequency (in kHz) + 433 500 400 RDFS(k) 300 200 100 0 0 0.2 0.4 0.6 0.8 1 1.2 1.4 Dim m ing Frequency (kHz)
Figure 6. RDFS vs. Dimming Frequency in LF Mode
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Micrel, Inc. The input frequency to the DRC pin can be 100Hz to 40kHz and the internal dimming frequency DR will be determined by RDFS. The duty cycle of the input frequency at DRC is converted according to Table 2 for the actual dimming duty cycle. For direct dimming control, connect DFS to VDD and use the MODE pin for the input dimming pulse. This method by passes the internal dimming control and allows for dimming control by the external PWM. DFS Filter In addition to the RDFS resistor-to-ground at the DFS pin, a series RC filter is required when operating at dimming frequencies below 1kHz. The reason is that the DFS pin is the output of a transconductance differential amplifier. The differential amplifier has a high-frequency pole. At low dimming frequencies of around 1kHz RDFS is high around 100k and the differential amplifier pole produces a phase shift that can cause instabilities in the DFS control. Therefore, a RC filter is required to compensate for the lagging phase shift created by the pole by adding a zero and therefore, a phase lead at the DFS pin. Use a 4k resistor in series with a 2.2nF ceramic capacitor. When using a dimming frequency of 2 kHz or less. The filter has no ill effect at higher dimming frequencies. COMP Connect a capacitor and resistor to ground to compensate the boost stage. DRC Dimming Ratio Control (DRC) is an input PWM dimming control. The MIC3263 converts this to one of sixteen dimming ratios that is used to dim the LEDs. The dimming ratio is built on a log scale. CINT CINT integrates the DRC input pulse. For a PWM frequency range of around 1kHz use 100nF. For a PWM frequency range of around 20kHz pulse, use 10nF. For a PWM frequency range of around 100Hz pulse use 1F. ISET Set the LED current of all six channels by this resistor. Use 2k for 30mA and 3k for 20mA. The RISET is inversely proportional to ILED. Use the following equation to find RISET:
60 RISET = ILED
MIC3263 For the best current matching accuracy design for an ILED current of 15mA to 30mA. CRV Use a 2.2F capacitor at the CRV pin. This is used as the reference voltage of the boost stage. The CRV capacitor is continually being charged or discharged in order to keep VOUT at the right level (refer to Functional Diagram illustration). CRV will be charged to keep the IO's at about 1.2V. IO1IO6 These are the connections to the linear-mode current amplifier in each channel. Connect the cathode end of the LED channels to these pins. The control loop will keep this at about 1.2V. 1.2V insures that the current amplifier is in the linear region and therefore can regulate the LED current. In cases where there are a different number of LEDs in a channel, the V(IO) of the channel with the fewest LEDs will have a higher V(IO). V(IO) can be as high as 7.5V before the fault monitoring circuits will sense that channel as a short to VOUT. When there are a different number of LEDs in a channel the IO voltage will be higher in the channels that have less LEDs in order to keep the LEDs biased correctly. A difference of up to 7.5V between channels can occur because of this. If the circuits that monitors the IO pins sees a fault, that channel will turn off and that channel's IO pin will be at high impedance. An off channel's IO pin will be near or below the booster output voltage. On a channel that has a shorted LED, that channel's IO voltage will increase to keep correct voltage drops on the other series LEDs. It is best to use equal number of LEDs in each channel but there will always be differences in the LEDs voltage drops so all IOs will not have the exact same voltage. Each channel has its own monitoring circuit monitoring the IO1IO6 pins. If any V(IO) drops below 0.5V (if an LED opens), that channel is turned off and the other channels are unaffected. If any IO goes about 7.5V (if several LEDs short to VOUT), that channel is turned off and the other channels are unaffected. VSW This is the boost-stage switch node, the collector of the internal power switch. EN Connect EN high to enable the part, low to disable. Do not leave the EN pin floating. VIN Supply voltage to the part (6V-40V).
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Micrel, Inc. VDD This is the output of the internal LDO regulator. Connect a 10F ceramic capacitor to this pin. PWM Dimming The duty cycle of the PWM pulse applied to the DRC input is converted to 16 log levels. This logarithmic dimming is a unique feature of the MIC3263 which better matches the sensitivity of the human eye compared to linear dimming. The DRC duty-cycle to DR duty-cycle conversion is shown in Table 2.
DRC Duty Cycle % PWM Dimming Ratio (DR) (N 1) / 7 DR = 10 - %
MIC3263
Figure 7. Duty-Cycle Thresholds and Hysteresis
N
0 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15
0 6.25 12.5 18.75 25 31.25 37.5 43.75 50 56.25 62.5 68.75 75 81.25 87.5 93.75
0 1.0 1.4 1.9 2.7 3.7 5.2 7.2 10 14 19 27 37 52 72 100
PWM Dimming Limits The minimum pulse width of the PWM Dim is determined by the PWM Dimming frequency and the L and C used in the boost stages output filter. At low-PWM Dimming frequencies, higher dimming ratios can be achieved:
T Dim Ratio = PWMD T LEDON
Figure 8. PWM Dimming Ratio
Table 2. Dimming Ratio
To avoid skipping between dimming levels, the MIC3263 uses Flicker-Free Dimming control. This technique uses a digital filter and hysteresis on the DRC pulse to provide a clean DR output. The digital filter has a 0.1F capacitor on the CINT pin to average the duty cycles of the PWM pulses. The averaged duty cycle has to be 4.16% higher than the nominal value before moving to the next dimming level as shown in Figure 7. Likewise, to move the previous dimming level the duty cycle has to be -4.16% lower than the nominal. To prevent flicker the duty-cycle hysteresis is set a 2%.
Consider that the human eye will perceive light flicker at a PWM dimming frequency below 100Hz. At 100Hz the time between pulses is 10s. If the PWM dimming minimum pulse width is 5s, then:
10ms 5s
Dim Ratio =
= 2000/1
If high dimming ratios are required, a lower dimming frequency is required. During each DR pulse, the inductor current has to ramp up to it steady state value to generate the necessary boost output voltage in order for the full programmed LED current to flow in the LED channels. The smaller the inductance value the faster this time is and a narrower PWM dimming pulse can be achieved. But smaller inductance means higher ripple current.
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Micrel, Inc. Figure 9 shows the waveforms during PWM dimming pulses. The DRC duty cycle is 75% and therefore the dimming ratio (DR) is 37%. Ch1 is the switch node. Ch2 is the sum of all six ILED channels. Figure 9 shows the boost converter is OFF (not switching) between PWM dimming pulses.
MIC3263
Figure 9. PWM Dimming Pulses (Ch1 Switch Node; Ch2 is the ILED Total)
Figure 10. Direct Dimming Control
Direct Dimming For direct dimming control connect DFS to VDD and use the MODE pin for the dimming pulse. This method will bypass the internal dimming control and allows for dimming control by the external PWM Dimming pulse (see Figure 9).
Boost Stage
A current-mode control is easier to compensate than voltage mode control, thus allowing for a less complex control loop stability design. An error amplifier amplifies the difference between the feedback voltage and the voltage on the CRV capacitor. This amplified error signal is called the VCONTROL. A PWM comparator compares the output of the error amp (VCONTROL) to the sum of inductor current and slope compensation currents. When the current sums reach VCONTROL, the PWM pulse is terminated and the boost power switch is turned off. A portion of the energy stored in the inductor flows into the output capacitor.
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MIC3263 Slope Compensation The boost stage uses peak current mode and requires slope compensation. Slope compensation is required to maintain internal stability of the boost stage across all duty cycles and to prevent any unstable oscillations. The MIC3263 uses a combination of internal slope compensation and a additional slope compensation that is set by an external resistor, RSLP. The ability to set the proper slope compensation through the use of a single external component results in design flexibility. This slope compensation resistor, RSLP, can be calculated as follows:
VOUT(MAX) - L xFsw 8.64 x10-6 x V
Figure 11. Boost Stage
RSLP =
IN(MIN)
The operating duty cycle can be calculated using the equation provided below:
(VOUT - eff x VIN ) VOUT
D=
and D = 1 - D
where VIN(MAX) and VOUT(MAX) can be selected to system specifications. The lowest value of RSLP should be 15k. Calculate RSLP using the lowest VIN and maximum VOUT the system will operate. Example: For these operating conditions: VIN(MIN) = 12V, VOUT(MAX) = 32V, L = 22H, FSW = 1MHz
RSLP = 32V - 22Hx1Mhz = 96.5k 8.64 x10-6 x12V
Find L using the following equation:
L= VIN xD IL_PP xFsw
IL_PP is the inductor peak-to-peak ripple current. Use a IL_PP of 20% to 40% of the total load current. FSW is the boost switching frequency. Output Capacitor In a boost converter, to find the COUT for a given VOUT ripple use the following calculation:
ILEDtotal xD VRIPPLE x Fsw
Use the next highest standard value. Table 3 compiles and lists RSLP values for one set of operating conditions. Select RSLP for VIN_MIN and VO_MAX.
COUT =
VRIPPLE can usually be kept below 50mV: ILED_TOTAL = 6 x 30mA = 180mA In the MIC3263, the LED current in each channel is individually regulated by that channels current amplifier (linear current regulator). These current regulators are fast enough to follow the boost output voltage ripple and to keep the LED ripple currents much lower than COUT can filter the output ripple voltage.
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MIC3263 From the small signal block diagram the loop transfer function is:
22H RSLP
VIN = 12V, VOUT = 32V F(kHz) 8.2H RSLP 10H RSLP
400 500 600 700 800 900 1000 1100 1200 1300 1400 1500 1600 1700 1800
2.77E+05 2.69E+05 2.61E+05 2.53E+05 2.45E+05 2.37E+05 2.30E+05 2.22E+05 2.14E+05 2.06E+05 1.98E+05 1.90E+05 1.82E+05 1.74E+05 1.66E+05
2.70E+05 2.60E+05 2.51E+05 2.41E+05 2.31E+05 2.22E+05 2.12E+05 2.03E+05 1.93E+05 1.83E+05 1.74E+05 1.64E+05 1.54E+05 1.45E+05 1.35E+05
2.24E+05 2.03E+05 1.81E+05 1.60E+05 1.39E+05 1.18E+05 96451 75231 54012 32793 15000 15000 15000 15000 15000
Figure 13. Simplified Voltage Control Loop
Equation 2: T(s) = Gea(s) x GVC(s) x H(s) where:
VCRV VOUT
Table 3. RSLC Values
H(s) =
and
Boost Compensation Current-mode control simplifies the compensation. In current mode the double pole created by the output L and C is reduced to a single pole. The explanation for this is beyond the scope of this data sheet, but it can be thought because the inductor current becomes a constant current source and can't act to change phase.
Gea (s) = gm Z o II R COMP +

sCCOMP
1
Equation 3:
VOUT ( s ) VCONTROL ( s )
Gvc (s) =
sL 1 2 1 D'RLOAD D' RLOAD = 2 Ri 1+ sRLOADCOUT 2
Figure 12. MIC3263 Current-Mode Loop Diagram
where RLOAD = Ai = 20 RCS = 0.02
VOUT IOUT
and Ri = Ai x Rcs = 0.4 .
AI and RCS are quantities that are internal to the MIC3263. The equation for GVC(S) is a theoretical model and should give an approximate idea of where the poles and zeros are located.
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MIC3263
Equation 3 shows that s =
2 D' RLOAD L
is a right-half plane
Error Amp The error amp is a gm type and the gain - GEA(S) - is: Equation 5:
zero (fRHPZ):
Equation 4:
2 D' RLOAD 2L
Gea (s) = gm Z o II R COMP +

sCCOMP
1
RHP Zero fRHPZ =
The loop bandwidth should be about 1/10 of the fRHPZ to ensure stability. From Equation 3, it is shown that there is only the single pole due to RLOADCOUT. This greatly simplifies the compensation. One needs only to get a bode plot of the transfer function of the control to output GVC(S) with a network analyzer. To measure GVC(S), tie CRV to a DC voltage source. Tie CRV to the steady state voltage that CRV will operate usually between 1V and 2.4V. By connecting CRV to a constant DC voltage, this effectively opens the CRV control loop and allows the measurement of the boost control loop. GVC(S) can be calculated with a computer using the above equation. From the bode plot of GVC(S) find what the gain of GVC(s) is at 1/10 of fRHPZ or less. Next design the error amp gain GEA(s) so the loop gain at the cross over frequency T(fCO) is 0db where fCO =1/10 of fRHPZ or lower.
40 20 Gain (db) phase (deg) 0 -20 -40 -60 -80 -100 1.E+02
Gain Phase
gm = 0.056mA/V and ZO = 5M. The error amplifier zero is 1 f = . Set the fCO at the mid band Zero 2R C
COMP COMP
where GEA(fCO) = gm x RCOMP. At fZERO x 10 the phase boost is near its maximum.
Figure 15. Internal Error Amp and External Compensation
Midband Gain
Fzero
Example 1 Conditions: VIN = 12V, VOUT = 29V, IOUT = 0.18A, L = 22H, COUT = 4.7F RLOAD = VOUT/IOUT = 161. When VCRV = 1.8V, the fRHPZ is:
fRHPZ = 2 D' RLOAD 2L = 162kHz
1.E+03
1.E+04 Freq
1.E+05
1.E+06
Figure 16 shows a plot of:
Figure 14. Error Amp Transfer Function
Gvc (s) = VCONTROL ( s ) VOUT ( s )
sL 1 2 1 D'RLOAD D' RLOAD = 2 Ri 1+ sRLOADCOUT 2
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Micrel, Inc. This example illustrates the RHPZ at 162kHz. Figure 16 details the -90 phase shift due to the RHPZ.
MIC3263
100
26db
Therefore R4 = 15k. Next set the error amplifier's zero at about 5kHz. Therefore C2 = 2.2nF. The location of the fZERO affects the phase boost in the loop transfer function. If fZERO were closer to 16kHz the phase boost would be less and vise versa.
50 Gain (db) phase (deg)
40
0
Gain (db) phase (deg)
20 0 -20 -40 -60
Gain Fzero
Midband Gain
-50
-100
Gain Phase
-150
-200 1.E+02
-80
Phase
1.E+03
1.E+04 Freq
1.E+05
1.E+06
-100 1.E+02
1.E+03
1.E+04 Freq
1.E+05
1.E+06
Figure 16. Control-to-Output Gain (GVC)
The goal is to make the loop transfer function T(fCO) crossover well before the RHPZ.
fRHPZ or less; chose fco = 16kHz . 10 From the plot and or calculation, the magnitude of:---
Figure 17. Error Amp Gain and Phase (in Example 1)
Chose a fco =
100 80 60 40 20 0
Gain
Gvc (16kHz) = 26db
H(s) = 20Log
1.8V = -24db 29V
Gain (db) phase (deg)
-20
Phase
Fco=1.6kHz
From:
T(s) = Gea (s) * Gvc (s) * H(s)
-40 1.E+02 1.E+03 1.E+04 Freq 1.E+05 1.E+06
T(16kHz) = Gea (16kHz) + 26db - 24db = 0 Gea (16kHz) = -2db 0.8v/v
Figure 18. Loop Gain and Phase (in Example 1)
0.8 = gm Z o II R 4 gm * R 4
(
)
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Design Procedure for a LED Driver
Symbol Input VIN IIN Output LEDs Chs VF VIO VOUT ILED/ch IOUT POUT DIM IN FDIM OVP FSW eff VDIODE Parameter Minimum Nominal Maximum Units
Input Voltage Input Current Number of LEDs Number of Channels Forward Voltage of LED Voltage Drop at the IO Pin Output Voltage LED Current/Channel Output Power PWM Dimming Dimming Frequency (internal) Output Over-Voltage Protection Switching Frequency Efficiency Forward Drop of Schottky Diode
8
12
14
V
8/Channel 6 3.4 1.1 28 30
8/Channel 6 3.6 1.2 30 30
1 5
8/Channel 6 4.0 2 34 30 0.18 6.2 100 40
Channels V V mA A W % kHz V MHz % V
80
1 85 0.5
Design Example
In this example, a boost six-channel LED driver operating off a 12V input is being designed. This design has been created to drive six channels of eight LEDs/channels for a total of 48 LEDs. The LED current will be set at 30mA. One is designing for 80% minimum efficiency at a switching frequency of 1MHz. For 34V out: Let R2=150k,
V xR2 1.8V x150k R1= CRV = = 8.39k VOUT - VCRV 34V -1.8V
Let VCRV = 2.2V therefore: VCRV x R2 VOUT - VCRV 2.2V x 150k 34V - 2.2V
R1 =
=
= 10.4k
Use the closest standard value of 10.5k. Therefore: VOVP = 2.4* (R2+R2)/R1 = 40V
Select RISET for a Given ILED
RISET = 60 ILED = 60 = 2k 30mA
Therefore: VOVP= 2.4* (R2+R2)/R1=45V. 45V is too high, meaning VCRV has to operate at a higher voltage than 1.8V. The CRV loop will charge the CRV capacitor to the necessary voltage to regulate.
Use 2k for RISET (R9)
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Switching Frequency Set RFSW To find the value of RFSW use the following equation:
MIC3263
Inductor Selection First calculate the RMS input current (nominal, minimum, and maximum) for the system given the operating conditions listed in the design example table. The minimum value of the RMS input current is necessary to ensure proper operation. Using Equation 7, the following values have been calculated: Equation 7:
IIN_RMS(MAX) = IIN_RMS(NOM) = VOUT(MIN) x IOUT(MIN) eff x VIN(MAX) eff x VIN(NOM) VOUT(MAX) x IOUT(MAX) eff x VIN(MIN)
= 0.43A (RMS)
RFSW(k) 500 - 0.3 x fSW(kHz) RFSW(k) 500 - 0.3 x (1000) = 200k Use 200kHz for RFSW (R5).
Dimming Frequency Select Resistor RDFS FDIM is 5kHz therefore HF mode is used. Connect MODE to VDD. To find RDFS (R8) use the following equation:
VOUT(NOM) x IOUT(NOM)
= 0.53A (RMS)
RDFS(in k) = 432 - 20 x FDIM(in kHz) = 432 - 20 x 10 = 232(k) The input frequency to the DRC pin can be 100Hz to 40kHz and the internal dimming frequency DR will always be 5kHz. The duty cycle of the input frequency at DRC is converted according to Table 2 for the actual dimming duty cycle. Since the dimming frequency is high the filter R6 and C6 is not necessary. They may be used with no ill effect.
Operating Duty Cycle The operating duty cycle can be calculated using Equation 6. Equation 6:
IIN_RMS(MIN) =
= 0.9A (RMS)
IOUT is the same as ILED total Selecting the inductor current (peak-to-peak) IL_PP to be between 20% to 50% of IIN_RMS(max), in this case 40%, we obtain: IL_PP(MAX) = 0.40 x IIN_RMS(MAX) = 0.4 x 0.9 = 0.36APP There is a trade off between the inductor value and the minimum PWM dimming pulse. The larger the inductor, the longer the PWM dimming pulse time will be. Due to this, the percentage of the ripple current may be limited by the required PWM dimming pulse. Also, the internal current amplifiers will attenuate the LED ripple current by more than a magnitude. It is recommended to operate in the continuous conduction mode. The value of "L" in Equation 8 represents Continuous Conduction Mode.
Equation 8:
DNOM =
(V (V
OUT(NOM)
- eff x VIN(NOM) - eff x VIN(MAX)
)
VOUT(NOM)
DMAX =
OUT(MAX)
)
VOUT(MAX)
DMIN =
(V
OUT(MAX)
- eff x VIN(MAX)
)
VOUT(MAX)
L=
VIN x D IL_PP x FSW
Therefore DNOM = 66%, DMAX = 80% and DMIN = 58%.
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Micrel, Inc. Using the nominal values, one gets:
12V x 0.66 0.36A x 1MHz
MIC3263 A Coilcraft # DO3316P-223ML is used in this example. Its DCR is 85 m, ISAT =2.6A. PINDUCTOR(MAX) = 0.92 x 85 m = 67mW
Output Capacitor In this LED driver application, the ILED ripple current is a more important factor compared to that of the output ripple voltage (although the two are directly related). To find the COUT for a required ILED ripple use the following calculation: For an output ripple ILED(RIPPLE) = 20mA. Equation 12:
ILED(total) x D VRipple x Fsw
L=
= 22H
If not a standard value, use the next higher standard value. Select the standard inductor value of 22H. Going back and calculating the actual ripple current gives:
VIN(NOM) x DNOM L x FSW 12V x 0.66 22H x 1MHz
IL_PP =
=
= 0.36APP
The average input current is different than the RMS input current because of the ripple current. If the ripple current is low, then the average input current nearly equals the RMS input current. In the case where the average input current is different than the RMS, Equation 9 shows the following:
Equation 9:
IIN_AVE(MAX) =
COUT =
VRIPPLE can usually be kept below 50mV: ILED(total) = 6 x 30mA = 180mA
(IIN_RMS(MAX) )
(0.9)
2
2 (IIN_PP ) 12 2
2
COUT =
0.18A x 0.76 50mV x 1Mhz
= 2.7F
IIN_AVE(MAX) =
(0.36) 12
0.9A
The Maximum Peak input current IL_PK can found using Equation 10:
Equation 10:
Use 2.7F or higher. The amount that COUT will discharge depends upon the time between PWM Dimming pluses and the size of the output capacitor. At the next PWM Dimming pulse COUT has to be charged up to the full output voltage VOUT before the desired LED current flows.
Input Capacitor The input capacitor is shown in the Typical Application. For superior performance, ceramic capacitors should be used because of their low Equivalent Series Resistance (ESR). The input capacitor CIN ripple current is equal to the ripple in the inductor. The ripple voltage across the input capacitor, is the ESR of CIN times the inductor ripple. The input capacitor will also bypass the EMI generated by the converter as well as any voltage spikes generated by the inductance of the input line. For a required VIN(RIPPLE). Equation 13:
IIN_PP 8 x VIN(RIPPLE) x FSW
IL_PK(MAX) = IIN_AVE(MAX) + 0.5 xIL_PP(MAX) = 1.0A The saturation current (ISAT) at the highest operating temperature the inductor must be rated higher than this. The power dissipated in the inductor is:
Equation 11:
PINDUCTOR(max) = IIN_RMS(MAX)2 x DCR
CIN =
=
(0.36A )
8 x 50mV x 1MHz
= 0.8F
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Micrel, Inc. This is the minimum value that should be used. To protect the IC from inductive spikes or any overshoot, a larger value of input capacitance may be required. Use 2.2F or higher as a good safe min.
Rectifier Diode Selection A Schottky diode is best used here because of the lower forward voltage and the low reverse recovery time. The voltage stress on the diode is the maximum VOUT and therefore, a diode with a higher rating than maximum VOUT should be used. An 80% de-rating is recommended here as well. Equation 14: Equation 17:
MIC3263
PWR SW_ON( MAX) = ISW_RMS(MAX) x VCE_ON_RMS (MAX)
ISW_RMS(MAX) = D(MAX) x IIN_AVE(MAX)2 +
D(MAX) x IIN_AVE(MAX)
(IIN_PP )2
12

VCE_ON_RMS (MAX) = D(MAX) x VCE_ON( MAX) PWR SW_ON( MAX) = D(MAX) x IAVE(MAX) x VCE_ON (MAX) PWR SW_ON( MAX) = 0.8 x 0.9A x 0.5V = 0.36W
Equation 18:
PWR SW_SWITCHING (MAX) = VOUT(MAX) x IIN_AVE(MAX) x tsw x Fsw
IDIODE_(MAX) = IOUT(MAX) = 0.18A
Equation 15:
PDIODE(MAX) VDIODE x IDIODE_(MAX) A SK34A is used in this example, it's VDIODE is 0.5V. PDIODE(MAX) 0.5V x 0.18A 0.09W
MIC3263 Power Losses To find the power losses in the MIC3263: There is about 25mA to 35mA input from VIN into the VDD pin. The internal bipolar power switch has an VCE(ON MAX) of about 0.5V.
tsw 20ns is the internal power switch on an off transition time
PWRSW_SWITCHING (MAX) = 34V x0.9 x 20ns x1MHz = 0.61W
Therefore: PMIC3263(MAX) = 14V x 35mA + 0.97 = 1.46W
Snubber If a high-frequency ringing is present at VSW, a snubber may be needed. A snubber is a damping resistor in series with a DC blocking capacitor in parallel with the power switch. When the power switch turns off, the drain to source capacitance and parasitic inductance will cause a high frequency ringing at the switch node. A snubber circuit as shown in the application schematics may be required if ringing is present at the switch node. A critically damped circuit at the switch node is where R equals the characteristic impedance of the switch node. Equation 18:
VCE(ON MAX) 0.5V
Equation 16:
PMIC3263(MAX) = VIN(MAX) x 35mA + PWRSW(MAX)
Where PWRSW(MAX) is the power loss of the internal bipolar power switch. The power switch power losses are the sum of the on-time losses; PWRSW(MAX) and the switching losses: PWRSW(SWITCHING MAX). PWRSW(MAX) = PWRSW(MAX) + PWRSW(SWITCHING MAX)
R SNUBBER =
LPARISITIC CDS
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Micrel, Inc. The explanation of the method to find the best R snubber is beyond the scope of this data sheet. Use RSNUBBER 2 1/2 W and CSNUBBER 470pF to 1000pF. If a snubber is used, the power dissipation in the RSNUBBER is: RSNUBBER = CSNUBBER x VOUT2 x FSW PSNUBBER = 470pF x 34V2 x 1MHz = 0.54W Table 2 illustrates the power losses in the Design Example.
Description Value
MIC3263
OVP The output voltage that the OVP will trigger is set according to Equation 19. Using the values for this example gives a max output voltage of: Equation 19:
VOVP= 2.4x (R2+R2)/R1=40V
RSLP To find RSLP use Equation 1 (which is repeated here): Use the minimum VIN and the maximum VOUT.
VOUT(MAX) - L x Fsw -6 8.64 x 10 x VIN(MIN)
Power Loss in the L Power Loss in the Schottky Diode MIC3263 Power Loss Maximum Total Losses Minimum Efficiency
0.069W 0.09W 1.46W 1.62W 80%
RSLP =
In this example:
34 - 22H x 1Mhz = 174k -6 8.64 x 10 x 8
Table 2. Major Power Losses
RSLP =
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Evaluation Board Schematic
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Bill of Materials
Item Part Number Manufacturer
(1)
Description
Qty.
C1 0603ZC222KAT2A C2, C6 C1608X7R1H222K GRM188R71H222K C3, C8 C4 C5750X7R1H106M 22205C106KAZ2A GRM21BR71A106KE51L 0805ZD106KAT2A 0603YC104KAT2A C5 C1608X7R1C104K GRM188R71C104K 0603ZD225KAT2A C7 GRM188R61A225KE34D C1608X5R1A225K D1 L1 R1 R2 R3 R4 R6 R5 R7 R8 R9
U1
Notes: 1. AVX: www.avx.com. 2. TDK: www.tdk.com. 3. Murata Tel: www.murata.com. 4. MCC: www.mccsemi.com. 5. Diode, Inc.: www.diodes.com. 6. Coilcraft: www.coilcraft.com. 7. Vishay: www.vishay.com. 8. Micrel, Inc.: www.micrel.com.
OPEN AVX TDK(2) muRata
(3)
2200pF, 10V, X7R, 0603
2
TDK(2) AVX(1) muRata(3) AVX(1) AVX(1) TDK(2) muRata
(3)
10F, 50V, X7R, 2220 10F, 10V, 0805
2 1
0.1F, 16V, X7R, 0603
1
AVX(1) muRata(3) TDK
(2)
2.2F, 10V, X5R, 0603
1
SK34A B349LA-13 DO3316P-223ML CRCW0603150KFKEA CRCW060310K0FKEA CRCW0603110KKFKEA CRCW060315K0FKEA CRCW060340K2FKEA CRCW0603200KFKEA CRCW0603100KFKEA. CRCW060326K7FKEA CRCW06032K00FKEA. MIC3263YML
MCC(4) Diode, Inc. (5) Coilcraft(6) Vishay Dale Vishay Dale
(7) (7)
Schottky 3A, 40V (SMA) 22H, 2.6A 150k 10k 110k (RSLP) 15.0k, 0603 (RCOMP) 4.02k 200k 100k 97.6k 2k
Six-Channel WLED Driver for Backlighting Applications
1 1 2 1 1 1 1 1 1 1
1
Vishay Dale(7) Vishay Dale(7) Vishay Dale Vishay Dale
(7) (7)
Vishay Dale(7) Vishay Dale Vishay Dale
(7) (7)
Micrel, Inc.(8)
January 2010
29
M9999-012110
Micrel, Inc.
MIC3263
Evaluation Board PCB Layout
January 2010
30
M9999-012110
Micrel, Inc.
MIC3263
Package Information
24-Pin 4mm x 4mm (MLF(R))
January 2010
31
M9999-012110
Micrel, Inc.
MIC3263
Recommended Land Pattern
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser's use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser's own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. (c) 2010 Micrel, Incorporated.
January 2010
32
M9999-012110

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